Synchronizing pulse generating method and method of receiving OFDM signal

ABSTRACT

In an OFDM receiver, a synchronisation pulse for defining a Fast Fourier Transform window is generated by examining the output of a correlator to find a sub-interval within which there is maximum correlation between samples of the symbol separated by the length of the useful part of the symbol. The synchronisation pulse is generated during this sub-interval ( 102 ). Adjustments to the timing of the synchronisation pulse are made only if the current error is significant and persistent. A signal representing the amount of adjustment is used to determine phase rotations applied to the output of the FFT circuit.

TECHNICAL FIELD

This invention relates to OFDM modulation. It is particularly concernedwith the generation of a synchronisation pulse representing an OFDMsymbol boundary, for example for use in Fourier Transform demodulation.

BACKGROUND ART

OFDM systems are well known. Various techniques have been used forsynchronisation of OFDM receivers. Some of these techniques requiretransmission of a special synchronisation signal. Other techniques relyon a standard OFDM signal, in which a complete symbol comprises a“useful part” and a “guard space”, the guard space sometimes beingreferred to as a guard interval, cyclic extension or cyclic prefix.

The guard space precedes the useful part of the symbol and contains arepeat of the data at the end of the useful part. (This is equivalent tohaving a guard space after the useful part, containing data which is thesame as that at the beginning of the useful part.)

Synchronisation techniques which rely upon the duplicated data in theguard space generally operate by performing a cross correlation betweencomplex samples spaced apart by the length of the useful part of thesymbol. This generates a timing pulse which is used in FourierTransformation of the received signal. The timing of the pulse is suchthat the Fourier Transform window contains only data from a singlesymbol.

If the timing is incorrect, inter-symbol-interference (ISI) occurs.However, the use of the guard space allows a certain amount of variationin the timing of the pulse while still avoiding ISI. The guard spaceshould be longer than the longest expected spread of delays amongstsignals received via different paths. The guard space is relativelysmall compared with the useful part of the signal; typically, the guardspace may contain Nu/32, Nu/16, Nu/8 or Nu/4 samples, where Nu is thenumber of samples in the useful part of the symbol.

Various techniques exist for deriving the synchronisation pulse from thecross-correlation. Although these operate adequately in common receptionconditions, there are circumstances in which the timing pulse isgenerated at an undesirable point, leading to ISI.

The cross-correlator, in the absence of noise or multi-pathinterference, produces an output which averages to zero except duringthe time that the guard space samples are cross correlated with thesamples, in the useful part of the symbol, which are of equal value.During that period, the cross-correlator produces a high-level output.This high-level output terminates at the end of one symbol and thebeginning of the next. One prior art arrangement integrates the outputof the correlator, and then peak-detects the resultant signal to producea timing pulse at the end of each symbol.

In the case of multi-path interference, wherein the same signal isreceived via different delays, in order to avoid ISI, thesynchronisation pulse should be generated during a window which has awidth equal to the overlap between the guard spaces of the two receivedsignals. However, the cross-correlator will produce a significant outputthroughout the period in which samples of either one or both of theguard space samples are being processed by the cross-correlator. In somecircumstances, this will result in the timing pulse being providedoutside the optimum window, thus resulting in ISI.

EP-A-0 772 332 describes other techniques for generating asynchronisation pulse. One such disclosed technique relies upon feedingthe output of the correlator to a phase locked loop (PLL). This can alsoresult in the synchronisation pulse being generated outside the optimumwindow in the case of significant noise or multi-path interference.Furthermore, a PLL requires a substantial number of symbol periods inorder to achieve lock, which therefore results in substantialacquisition time.

A further problem which can arise in prior art arrangements results fromthe fact that, when the synchronisation pulse is adjusted as a resultof, for example, changing signal conditions, the complex values in thefrequency bins at the output of the FFT suffer varying degrees of phaserotation. Although a subsequent channel estimator and corrector canhandle these changes, this can result in a further increase ofacquisition time and requires a significant amount of processing power.

It would therefore be desirable to provide a technique for generating asynchronisation pulse in which these problems are avoided or at leastmitigated.

DISCLOSURE OF INVENTION

Aspects of the present invention are set out in the accompanying claims.

Namely, this invention is a method of generating a synchronisation pulserepresenting a symbol boundary in an OFDM signal comprising usefulsymbol periods separated by guard spaces, with data in each guard spacecorresponding to part of the data in a respective useful period, themethod comprising a step of providing a signal representing the degreeof correlation between samples of a received signal which are separatedby a period corresponding to the useful part of the symbol, the signalthus providing an output representing for each symbol an interval duringwhich significant correlation is found, and a step of determining asub-interval within which a maximum degree of correlation occurs andarranging for the synchronisation pulse to be provided within thissub-interval.

According to a further aspect, a synchronisation pulse is generated byproviding a signal representing the degree of correlation betweensamples of a received signal which are separated by a periodcorresponding to the useful part of the symbol, the signal thusproviding an output representing an interval during which significantcorrelation is found, the method comprising the step of determining asub-interval within which a maximum degree of correlation occurs, andarranging for the synchronisation pulse to be provided within thissub-interval.

In the case of multi-path interference, it is found that the degree ofcorrelation is at a maximum throughout a period whose width correspondsto the overlap of the guard spaces. This is an optimum period forgeneration of the synchronisation pulse, because this will ensure thateach Fourier Transform window contains samples from only one symbol,even though the same symbol is received with different delays. Using thetechniques of the present invention, the output of the cross-correlatoris examined to determine when this optimum period occurs.

In a preferred embodiment, the output of the cross-correlator iscompared with a threshold, and the optimum sub-interval defining theperiod in which the synchronisation pulse is to be generated isrepresented by the period during which the cross-correlator outputexceeds this threshold. Preferably, the threshold is varied independence upon the output of the cross-correlator, and more preferablythe threshold is based upon the maximum level of the correlator output.

The use of a threshold is regarded to be an independently inventiveaspect of the invention. According to this further aspect, the output ofa correlator which represents the degree of correlation between samplesof a received signal which are separated by a predetermined number ofsample intervals is applied to a level detector, and only those parts ofthe signal which exceed a predetermined (preferably variable) level aretaken into account in determining the time-at which a synchronisationpulse should be generated.

If desired, the timing pulse could be generated at any time during thewindow which represents maximum correlation, for example at the middleof this window. As the signal conditions vary, this point may shift, inwhich case the synchronisation pulse will alter accordingly. In thepreferred embodiment, however, the timing of the synchronisation pulseis altered only if certain conditions are met. For example, the timingcan be altered only if the current timing is found to be in error apredetermined number of times, and/or only if the current error exceedsa predetermined amount. This technique, which is regarded to be anindependently inventive aspect, avoids excessive numbers of changes inthe timing of the Fourier Transform operation, each of which would causea phase rotation of each of the carriers at the output of the FFT by adifferent angle, which would place a heavy workload on the channelestimator conventionally provided.

According to a still further aspect of the invention, there is providedat the output of the FFT means for imparting different phase rotationsto the respective samples of the FFT output, this means being responsiveto a signal representing the amount of shift imparted to asynchronisation pulse by a pulse generator for determining the amount ofphase rotation applied. This allows very rapid, and indeed possiblyinstantaneous, compensation for changes in the timing of thesynchronisation pulse. The phase rotations may be imparted by a circuitpositioned between the FFT and the channel estimator and corrector, oralternatively the channel estimator and corrector can perform the phaserotation. Preferably, changes in the timing of the synchronisation pulseare arranged to occur relatively infrequently (according to the aspectof the invention mentioned previously), and preferably only, ornormally, in predefined amounts. This facilitates the determination ofthe appropriate phase rotations to be applied to the FFT outputs. Thesephase rotations can be calculated in response to a signal representingan actual or prospective degree of shift in the timing of thesynchronisation pulse, or alternatively could be derived from a look-uptable addressed in accordance with such a signal.

In the prior art mentioned above, the output of the correlator isfiltered, for example by using a sliding-window averager which sums themost recent Ng samples from the cross-correlator. This filteringtechnique would, however, vary the shape of the correlator output, andaccordingly, in the preferred embodiment of the present invention, thecross-correlator output is filtered by summing the most recent L1samples, where L1 is significantly smaller than Ng.

It is known in the prior art to take the output of the sliding-windowaverager and process this to provide a signal used in fine frequencycorrection. This technique is preferably also used in arrangementsaccording to the present invention. However, to obtain better quality offine frequency estimation, in the preferred embodiment of the presentinvention, a second filter is applied to the output of the first filter,and produces an output which represents averaging over a number ofsamples which is substantially greater than L1. For example, the outputmay be equivalent to what would be obtained from a single filter summingthe latest Ng samples.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an OFDM receiver in accordance with thepresent invention;

FIG. 2 schematically represents an OFDM signal;

FIG. 3 is a block diagram of a prior art arrangement for generating asynchronisation pulse;

FIG. 4 schematically represents the effects of multi-path interferenceon the cross-correlation output;

FIG. 5 is a block diagram of a synchronisation circuit of the presentinvention;

FIG. 6 is a block diagram of a timing recovery circuit forming part ofthe synchronisation circuit;

FIG. 7 represents part of a typical waveform derived from thecorrelation output, this part occupying a sub-interval within which asynchronisation pulse can be optimally generated; and

FIG. 8 represents how the three values of Ns, L1 and L2 may vary.

BEST MODE FOR CARRYING OUT THE INVENTION

Referring to FIG. 1, an OFDM receiver 2 comprises an antenna 4 whichreceives a signal and presents it to a down-converter 6 which convertsthe RF signal to an IF signal. This is then converted into a basebandsignal by an IF-to-baseband converter 8. This produces at its outputcomplex samples of each transmitted OFDM symbol. These complex samplesare digitised by an analog-to-digital (A/D) converter 10, and deliveredvia a fine frequency adjustment circuit 12 to a Fast Fourier Transform(FFT) circuit 14. The FFT circuit 14 converts the samples from the timedomain to the frequency domain, and the symbol data at the output isprovided to a phase rotator 15, a channel estimator and corrector 16 anda decoder 17.

The techniques of the present invention facilitate the provision of afeed-forward system, which doesn't rely on feedback or phase lockedloops (PLL's) for adjusting the local oscillator frequencies. However,it would be possible in an alternative arrangement if desired to providesuch feedback, so that the synchronisation circuit 18 would beresponsive to, for example, the complex samples from the A/D converter10 and and/or a signal from the channel estimator and corrector 16.

The complex samples are delivered to a symbol synchronisation circuit20, which generates a frequency offset signal for the fine frequencyadjustment circuit 12, and a synchronisation pulse for use by the FastFourier Transform (FFT) circuit 14. The FFT circuit 14 requires thesynchronisation pulse so that each transformation operation is alignedwith the start of the OFDM symbol.

The circuit described so far is known from the prior art, with theexception of the phase rotator 15. The present invention is directed,inter alia, to a novel and inventive technique for use in the symbolsynchronisation circuit 20.

Referring to FIG. 2( a), it is assumed that an OFDM symbol consists ofNu+Ng samples, representing Nu samples in the useful part U of thesignal, preceded by Ng samples in the guard space G. The Ng samples inthe guard space G contain the same data as the last Ng samples of theuseful part U of the symbol (as indicated, in respect of one of thesymbols, by hatching).

Referring to FIG. 3, in a prior art synchronisation circuit, the complexsamples from the IF-to-baseband converter 8 are provided in successionto a first-in first-out (FIFO) register 30 of a cross-correlator 28.This register contains Nu stages, so that it provides a correspondingdelay of Nu samples. The output of the register 30 is provided to acomplex conjugator circuit 32 of the correlator 28, which converts eachsample into its complex conjugate. Then, at a multiplier 34 of thecorrelator 28, each complex conjugate is multiplied by an undelayedsample from the A/D converter 10. (Alternatively, the complex conjugator32 can be inserted into the other path to the multiplier 34.)

Whenever the complex conjugates of the delayed samples in the guardspace G are multiplied by the samples of equal value derived from theend of the succeeding useful part U of the symbol, the correlator outputwill be high. At other times, the correlator output will adopt a randomvalue. FIG. 2( b) represents the output of the correlator. For clarity,FIG. 2( b) represents an ideal output after averaging over a number ofsymbols, although in practice the averaging can occur at a later stage.

The output of the correlator 28 is provided to another FIFO register 36,this register containing Ng locations. An integrator 38 receives theoutput of the FIFO register 36 as well as, directly, the output of thecorrelator 28. The integrator serves to add each new sample to thecurrent integrator output and subtract the sample received Ng samplesearlier. Thus, the output represents the sum of the most recent Ngsamples. The output is represented at FIG. 2( c). It will be noted thatthis output gradually increases towards the end of each symbol, and thenimmediately starts decreasing. A peak detector (not shown) produces atiming signal whenever the integrator output reaches a peak (for exampleas shown at timing t in FIG. 2). This is used as the synchronisationpulse for the FFT 14, and it will be noted that it will appear at theend of each symbol, i.e. exactly when the most recent Nu samplesreceived in the FIFO register 36 are the appropriate ones for use by theFFT 14.

The FFT operates on the Nu samples of the useful part U of the signal.It will be appreciated from FIG. 2 that the synchronisation signal tcould be provided at any time within the last Ng samples of a symbol(i.e. whenever the FIG. 2( b) waveform is at a high level), and stillavoid ISI, because the provision of the guard space G means that thepreceding Nu samples will be from the same signal.

FIG. 4 shows one possible effect of multi-path interference. FIGS. 4( a)and 4(b) show the same signal, received at different times, FIG. 4( a)representing the weaker of the signals, which in this case is receivedfirst.

FIG. 4( c) represents the output which the correlator of FIG. 3 wouldprovide in the absence of the FIG. 4( b) signal, and FIG. 4( d) showsthe output which the correlator would provide in the absence of thesignal of FIG. 4( a). With both signals present, the correlator producesan output represented at FIG. 4( e). (Again, FIGS. 4( c) to 4(e)represent correlator outputs averaged over a plurality of symbols.)

The waveform in FIG. 4( e) has three sections, 101, 102 and 103. Thesesections collectively represent an interval during which the correlatoroutput is at a high-level due to significant correlation between valuesseparated by Nu samples in one or both of the signals. The highestcorrelation, in sub-interval 102, occurs when positive correlationsresult from both signals. It will be noted that section 102 is the onlypart of the waveform in which the last Ng samples of a symbol fromsignal 4(a) occur at the same time as the last Ng samples of a symbol ofsignal 4(b). Accordingly, sub-interval 2 is the only period in which atiming signal can be provided while avoiding ISI.

However, if the correlator output were to be integrated by the prior artcircuit of FIG. 3, the output would be represented by FIG. 4( f). Thepeak of this output occurs at the end of section 3, which means that itwould be too late. In particular, this means that although the FFT 14would process only samples from symbol i of the FIG. 4( b) signal, itwould additionally process samples from symbol i+1 of the signal of FIG.4( a).

Referring to FIG. 5, the synchronisation circuit 20 of the presentembodiment of the invention comprises a correlator 28 formed by shiftregister 30, complex conjugator 32 and multiplier 34, as in the priorart arrangement of FIG. 3. The output of the correlator 28 is deliveredto an averager 46, which can also be an FIFO register as in the priorart arrangement of FIG. 3, but in this case the number of stages isequal to L1, which is significantly less than Ng. The output of the FIFO46 is delivered to a symbol averager 48, which sums each sample fromFIFO 46 with the corresponding samples from the preceding Ns symbols.Accordingly, the output of the symbol averager is equal to thecorrelator output averaged over L1 samples and Ns symbols. In the caseof multi-path interference as shown in FIG. 4, the output will besimilar to the waveform shown in FIG. 4( e), with slight smoothingresulting from the L1 averaging.

This output is then delivered to a timing recovery circuit 50. Thisprovides the synchronisation signal to the FFT 14.

The functions performed by the timing recovery circuit 50 areillustrated schematically by the blocks in FIG. 6. The output samplesfrom the symbol averager 48 are delivered to an absolute value circuit52. This calculates the absolute value for each sample, i.e. √(x²+y²),where x and y are the in-phase and quadrature components of the sample.These are checked in a peak detector 54 which determines the value ofthe sample with the largest magnitude. A window generating circuit 56responds to the samples from the absolute value circuit 52 and the peakvalue detected by the circuit 54 to determine the nearest samples oneither side of the peak that are below a threshold which is equal to0.75 times the peak value. The window generator 56 will therefore detecta range of samples, from n_(min) to n_(max) which represents the largestdegree of correlation in the signal from the correlator. FIG. 7represents a typical waveform representing the samples from the symbolaverager 48 during this period. Timing signals generated during theperiod n_(min) to n_(max) are likely to be suitable for avoiding ISI.

A synchronisation signal generator 58 generates a synchronisation pulseonce per symbol, following an initialisation operation described below.

A comparator 60 compares the time at which this timing signal isgenerated with the range of sample values n_(min) to n_(max) determinedby the window generator 56. If there is a significant difference, avalue stored in a counter circuit 62 is altered. When one of severalvalues stored in the counter circuit 62 reaches a predeterminedthreshold, a signal is sent to the signal generator 58 to adjust thetiming of the synchronisation signal by an amount which depends upon therange n_(min) to n_(max) calculated by the window generator 56. Thearrangement is such that the timing signal will tend to be generatedabout midway between the samples n_(min) and n_(max), but that it willonly be adjusted if there are persistent and/or significant errors inthe current timing signal.

In this embodiment, the comparator 60 divides the range n_(min) ton_(max) into four quarters, q1, q2, q3 and q4, in order of increasingsample number, as shown in FIG. 7. If the comparator 60 determines thatthe current timing of the synchronisation pulse lies in q1, then afirst, “early” register in counter 62 is incremented by one. If thetiming signal is found to lie in q4, then a second, “late” register isincremented by one. If the timing lies in q2 or q3, then both countersare decremented by one, although they are not allowed to go below zero.If at any time either counter reaches the value 4, then the countercircuit 62 causes the timing pulse generated by signal generator 58 tobe shifted by an amount corresponding to (n_(max)−n_(min))/4 rounded tothe nearest four samples, for the next symbol (or for a predeterminedlater symbol, for example the second or third succeeding symbol, toallow more time for the further processing described below). The timingpulse is moved forward or backwards depending upon whether it is theearly or late counter which has reached the value 4.

A further register of counter circuit 62 is incremented or decrementeddepending on whether the timing lies outside the range n_(min) ton_(max). If this occurs over four successive periods, the countercircuit 62 causes an initialisation operation.

This initialisation operation, which would occur when re-tuning to a newstation or after power-on, results in the signal generator 58 being setto generate the timing signal at a position midway between n_(min) andn_(max). The initialisation operation also involves effecting changes tofilter characteristics, as described below.

Whenever the signal generator 58 is caused to shift the timing of thesynchronisation pulse, this will result in differential phase rotationof the carriers at the output of the FFT 14. To facilitate the handlingof this, the counter circuit 62 of the timing recovery circuit 50outputs a signal representing the amount of change applied to thesynchronisation pulse, which signal is received by the phase rotator 15.The phase rotator 15 contains a look-up table storing pre-computed phaserotations corresponding to the possible values represented by the signalfrom the timing recovery circuit 50. Accordingly, upon receipt of thissignal, the appropriate values are derived from the look-up table andthe respective complex samples in the FFT output are adjusted bycorresponding amounts. As an alternative, the phase rotator 15 can havemeans for computing the phase rotations in response to the signal fromthe timing recovery circuit 50. It will be noted that phase adjustmentis facilitated because:

-   -   (a) the timing recovery circuit 50 produces a signal        representative of the amount of adjustment of the        synchronisation pulse;    -   (b) the timing recovery circuit is arranged, as described above,        such that adjustments occur relatively infrequently;    -   (c) the amount of adjustment of the synchronisation pulse is        rounded, which reduces the number of different possible        adjustments applied to the timing of the synchronisation pulse;    -   (d) the timing recovery circuit is capable of specifying in        advance the first symbol to be affected by the change in timing        of the synchronisation pulse;    -   (e) because timing adjustment is performed only after the timing        recovery circuit 50 has detected a succession of timing errors        of a similar nature, it is possible if desired for the        determination of appropriate phase rotations to be carried out        in advance, for example when only one or two symbols have been        determined to have timing errors, to allow even more time for        the operation; and    -   (f) the changes to be applied can be pre-computed and stored in        the look-up table.

Referring again to FIG. 5, the output of the FIFO register 46 is alsodelivered to a further FIFO register 64, which acts as a sliding-windowaverager summing successive groups of L2 samples. The sampling rate isdivided by L1, and the L2 most recent samples are summed. Preferably,L1×L2 is substantially equal to Ng. The combination of these twoaveragers, 46 and 64, is functionally equivalent to the conventionalaverager 36 in the prior art circuit of FIG. 3. The output of theaverager 64 is presented to a peak search circuit 66, which finds thesample with the largest magnitude and derives the angle of this sample,which provides an estimate of the fine frequency deviation. A signalrepresenting this frequency offset is then provided to the frequencycorrection circuit 12 which corrects the frequency by phase rotation ofthe received samples.

In this embodiment, the L2 averager 66 averages successive values withina symbol, but alternatively the averaging could be performed overcorresponding values in successive symbols (although this would delaythe provision of an accurate fine frequency estimate).

Upon powering-up of the receiver, or when tuning to a new station, it isdesirable to lock on to a new signal as soon as possible. This processpreferably begins with the first received symbol. In this case, thevalue Ns, i.e. the number of symbols taken into account by the symbolaverager 48, would start at 1, and would therefore increase for eachnewly-received symbol, although preferably the value would not beallowed to increase beyond a relatively small number (e.g. 8) so as toavoid too long a period, during which the signal may change, being takeninto consideration.

Because Ns starts at a very low value and then increases, it isdesirable to have the values L1 and L2 vary during this initial stage.L1 preferably starts at a relatively high value (although stillpreferably substantially smaller than Ng) because otherwise with lowvalues of Ns the output of the L1 averager would be likely to beexcessively erratic. Setting L1 equal to, for example, 64 while Nsequals one would provide a good first timing estimate for thesynchronisation signal from the first symbol. If L1 is initially setrelatively high, then preferably L2 is set correspondingly low tocompensate.

A table of FIG. 8 represents an example of how these values may vary.

The values for the 9th and succeeding symbols remain at the values forthe 8th symbol.

It will be appreciated that the present invention is effective not onlyin simple multi-path interference such as that described in connectionwith FIG. 4, but also in other situations where signals are received viamore than two paths. In these circumstances, the waveform of FIG. 4( e)would be a more complex staircase waveform. However, so long as thespread of delay is such that there is a period in which all the guardspaces overlap, the techniques of the invention can be used to determinea corresponding window in which the synchronisation signal should begenerated.

It is to be noted that, although reference has been made to the periodin which there is overlap of the guard spaces, which are at thebeginning of the symbols, as in the embodiment described above this isnot necessarily the correct time to generate the synchronisation signal;in the embodiment above, there is a corresponding period, in the overlapof the duplicated data, when the signal should be issued. The choice ofthe appropriate interval will depend upon a number of factors, such aswhether the guard space is considered to be at the beginning or the endof the signal, and whether (as in the above embodiment) the timingsignal is used to define the end of a symbol period, rather than thebeginning. It is to be further noted that the description set out abovedisregards the delays which may occur, for example in the FIFO averager46. In the embodiment described above, in practice, it is appropriate tomake a correction corresponding to −(L1)/2 samples to account for thisdelay.

The above embodiment correlates samples spaced by Nu sample periods bymeans of multiplying one sample with the complex conjugate of the othersample. Other arrangements are possible. For example, the correlatorcould operate by taking the difference between the absolute values ofsamples separated by Nu sample periods, as described in our co-pendingUK patent application no. BPA9920446.3(agent's reference J00041703 GB).

The invention has been described in the context of an OFDM receiver, inwhich the synchronisation pulse is used to define the window of sampleson which a Fast Fourier Transformation is performed. However, theinvention is also useful in other circumstances in which there is a needfor a synchronisation pulse representing the symbol boundaries; forexample, such a pulse would be valuable in a repeater where full FFTdemodulation is not performed.

The functional elements described herein can be implemented either indedicated hardware or in software.

INDUSTRIAL APPLICABILITY

As has been described, the method of generating a synchronisation pulserepresenting a symbol boundary in an OFDM signal according to thisinvention comprising useful symbol periods separated by guard spaces,with data in each guard space corresponding to part of the data in arespective useful period, the method comprising a step of providing asignal representing the degree of correlation between samples of areceived signal which are separated by a period corresponding to theuseful part of the symbol, the signal thus providing an outputrepresenting for each symbol an interval during which significantcorrelation is found, and a step of determining a sub-interval withinwhich a maximum degree of correlation occurs and arranging for thesynchronisation pulse to be provided within this sub-interval.Therefore, the processing time including acquisition time can beshortened and the processing power can be decreased while still avoidingthe inter-symbol-interference (ISI).

1. A method of generating a synchronisation pulse representing a symbolboundary in an OFDM signal comprising useful symbol periods separated byguard spaces, with data in each guard space corresponding to part of thedata in a respective useful period, the method comprising providing asignal representing the degree of correlation between samples of areceived signal which are separated by a period corresponding to theuseful part of the symbol, the signal thus providing an outputrepresenting for each symbol an interval during which significantcorrelation is found, the method comprising determining respectivedegrees of correlation in each of plural sub-intervals within saidinterval, detecting a sub-interval within which a maximum degree ofcorrelation occurs, and providing a synchronisation pulse within thedetected sub-interval.
 2. A method as claimed in claim 1, wherein thedetected sub-interval is determined by applying a threshold to thesignal representing the degree of correlation.
 3. A method as claimed inclaim 2, wherein the threshold is varied.
 4. A method as claimed inclaim 3, wherein the threshold represents a value which is dependentupon the maximum value of the signal representing the degree ofcorrelation.
 5. A method as claimed in claim 1, in which the signalrepresenting the degree of correlation is subject to filtering prior tousing the signal to determine said detected sub-interval, the filteringbeing such that each filtered output sample represents, substantially,an average of a predetermined number of successive samples, saidpredetermined number being substantially less than the number of sampleswithin a guard space.
 6. A method as claimed in claim 5, in which thefiltered output represents values averaged over a plurality of symbols.7. A method as claimed in claim 6, in which the number of symbols overwhich the filtered output values are averaged increases during anacquisition stage, and in which the filtering is adjusted during thatacquisition stage so as to decrease the number of successive samples,the average of which is represented by each filtered output sample.
 8. Amethod as claimed in claim 5, wherein the filtered output is subjectedto further filtering before being processed to provide a signalrepresenting a fine frequency offset.
 9. A method as claimed in claim 1,including the step of adjusting the timing of the synchronisation pulseonly if a calculated error in the current timing exceeds a predeterminedthreshold.
 10. A method as claimed in claim 1, including the step ofadjusting the timing of the synchronisation pulse only if the currenttiming is determined to be in error over a predetermined number ofsymbol periods, the predetermined number of symbol periods being greaterthan one.
 11. A method as claimed in claim 1, wherein the timing of thesynchronisation pulse is adjusted in predetermined quantitiescorresponding to a plurality of sample periods.
 12. A method ofgenerating a synchronisation pulse representing a symbol boundary in anOFDM signal comprising useful symbol periods separated by guard spaces,with data in each guard space corresponding to part of the data in arespective useful period, the method including: calculating an error inthe current timing; comparing the magnitude of the calculated error witha predetermined threshold; and adjusting the timing of thesynchronisation pulse when the magnitude of the calculated error in thecurrent timing exceeds said predetermined threshold.
 13. A method ofgenerating a synchronisation pulse representing a symbol boundary in anOFDM signal comprising useful symbol periods separated by guard spaces,with data in each guard space corresponding to part of the data in arespective useful period, the method including the step of (i) countingthe number of symbol periods over which the current timing is determinedto be in error, and (ii) adjusting the timing of the synchronisationpulse in response to the counted symbol periods exceeding apredetermined number greater than one, wherein the error in the currenttiming is calculated by comparing a timing signal with a range of valuescorresponding to a sub-interval, said sub-interval being detected bydetermining respective degrees of correlation in each of pluralsub-intervals within a correlation interval within which significantcorrelation occurs, and detecting said sub-interval within which amaximum degree of correlation occurs.
 14. A method as claimed in claim13, wherein the timing of the synchronisation pulse is adjusted inresponse to the current timing having an error exceeding a predeterminedthreshold over said predetermined number of symbol periods.
 15. A methodof generating a synchronisation pulse representing a symbol boundary inan OFDM signal, said signal comprising symbols, each symbol being formedof successive complex samples, each of said successive complex sampleshaving a sample period, and each symbol including useful symbol periods,said useful symbol periods being separated by guard spaces, with data ineach guard space corresponding to part of the data in a respectiveuseful symbol period, the method including: changing the timing of thesynchronisation pulse by a predetermined adjustment quantity, saidpredetermined adjustment quantity being calculated by a roundingoperation so that said predetermined adjustment quantity is equal to aplurality of sample periods.
 16. A method as claimed in claim 12,wherein the timing of the synchronisation pulse is adjusted inpredetermined quantities corresponding to a plurality of sample periods.17. A method of receiving an OFDM signal, the method including the stepof generating a synchronisation pulse using a method as claimed in anyone of claims 1, 12, 13 and 15, and using the synchronisation pulse inorder to apply a Fast Fourier Transform to complex samples derived fromthe OFDM signal.
 18. A method according to claim 17, the method furtherincluding the step of providing, when the timing of the synchronisationpulse is altered, a signal representing the degree of alteration, andapplying to the transformed samples phase rotations determined by thissignal.
 19. A method as claimed in claim 18, wherein the phase rotationsare determined by values in a look-up table addressed in accordance withthe signal representing the degree of alteration of the synchronisationpulse timing.
 20. A method of receiving an OFDM signal, the methodincluding: generating a synchronisation pulse and using thesynchronisation pulse in order to apply, in a FFT unit, a Fast FourierTransform to complex samples derived from the OFDM signal; providing,once the timing of the synchronisation pulse is altered, a signalrepresenting the degree of alteration to a phase rotation unit; andapplying, to the transformed samples received from the FFT, phaserotations determined based on the signal.
 21. A method as claimed inclaim 20, wherein the phase rotations are determined by values in alook-up table addressed in accordance with the signal representing thedegree of alteration of the synchronisation pulse timing.
 22. Apparatusfor generating a synchronising pulse, the apparatus operating accordingto a method as claimed in claim
 1. 23. An OFDM receiver arranged tooperate in accordance with a method according to claim
 17. 24. A methodof generating a synchronisation pulse representing a symbol boundary inan OFDM signal comprising useful symbol periods separated by guardspaces, with data in each guard space corresponding to part of the datain a respective useful period, the method including the steps of:adjusting the timing of the synchronisation pulse in predeterminedquantities corresponding to a plurality of sample periods, receiving anOFDM signal, by generating the synchronisation pulse, using thesynchronisation pulse in order to apply a Fast Fourier Transform tocomplex samples derived from the OFDM signal in a FFT unit, providing toa phase rotation unit, once the timing of the synchronisation pulse isaltered, a signal representing the degree of alteration, and applying,to the transformed samples received from the FFT unit, phase rotationsdetermined based on the signal.
 25. A method as claimed in claim 24,wherein the phase rotations are determined by values in a look-up tableaddressed in accordance with the signal representing the degree ofalteration of the synchronisation pulse timing.